Methods and apparatus for signal amplitude control systems

ABSTRACT

A method for obtaining a rectified signal from a first alternating current signal. The method includes the step of inputting the first alternating current signal into a variable gain amplifier to obtain a second alternating current signal. The second alternating current signal has a substantially constant peak-to-peak voltage irrespective of a power level of the first alternating current signal. The method further includes the step of rectifying the second alternating current signal, using a power detector circuit, to obtain the rectified signal, whereby a direct current level of the rectified signal is substantially proportional to the power level of the first alternating current irrespective of the power level of the first alternating current. The rectified signal may then be employed in, for example, a feedback control circuit to control the amount of power output by an RF signal source.

BACKGROUND OF THE INVENTION

The present invention relates to amplitude-controlled signal sources.More particularly, the invention relates to methods and apparatus forsignal amplitude control systems.

Radio frequency (RF) signal sources are circuits that generate, inresponse to a control input, amplitude-controlled output signals havingvariable amplitudes. FIG. 1 is a block diagram illustration showing anRF signal source 100, which receives a user-supplied control signal 102.User-supplied control signal 102 is a direct current (d.c.) voltagesignal and is typically derived from the digital output of amicroprocessor, a microcontroller, or similarly suitable logic circuitsthrough, for example, a conventional digital-to-analog converter (DAC).Responsive to the d.c. voltage level on user-supplied control signal102, RF signal source 100 generates an amplitude-controlled outputsignal 104. The amplitude-controlled output signal 104 may then beemployed to drive a load 106, representing a resistive load or anantenna such as that found in a cordless or a cellular phone of modemcommunication systems.

Ideally, all RF signal sources should produce the same output powerlevel for a given control input d.c. voltage level. Due totemperature-induced variations, manufacturing tolerances, and others,however, the parts within different RF signal sources may cause those RFsignal sources to output different levels of power for a given controlinput d.c. voltage level. To keep the output power of RF signal sourceswithin a specified range, a calibration scheme is usually necessary.

If the response of the RF signal source is linear, i.e., its outputpower level or voltage level varies linearly with the d.c. voltage levelof its control input, and there is no temperature-dependentfluctuations, calibration may be performed in a relativelystraight-forward manner. To facilitate discussion, FIG. 2 shows a line110 on a plot of RF signal source output power level (shown in dBm) vs.the d.c. voltage level of the RF signal source control input signal.Line 110 represents the ideal linear response of an RF signal source, inwhich a linear increase or decrease in the control input d.c. voltagelevel results in a corresponding linear increase or decrease in theoutput power level of the amplitude-controlled output signal.

The ideal RF signal source associated with line 110 may be calibrated bymeasuring its output power levels at two points on line 110, e.g.,points 112 and 114, and noting the required control input d.c. voltagelevels at those two points. The d.c. voltage level of the control inputthat is required to obtain any desired output power level on line 110,e.g., at point 116, may then be calculated simply by extrapolating alongthe slope of line 110. By appropriately modifying the control inputsignal voltage level, the ideal RF signal may then be compensated, e.g.,for its manufacturing tolerances.

Unfortunately, prior art RF signal sources have been far from ideal. Theresponse of a typical prior art RF signal source may, for example,resemble that of curve 120 due to the nonlinear relationship between itsoutput power vs. the d.c. voltage level of the control input. Because ofthis nonlinearity, the calibration of the prior art RF signal sourcesrequires much additional work. For example, it is not possible to derivethe required d.c. voltage level of the control input at point 122 simplyby calibrating at points 124 and 126 on curve 120. To calibrate theprior art RF signal source that is associated with curve 120, it istypically necessary to learn its response by sampling along line 120 atdiscrete intervals. From the sampling data, a lookup table may then beconstructed. This lookup table may then be employed to modify the d.c.voltage level of the control input (by, e.g., modifying theaforementioned digital signal from the microcontroller) to calibrate theprior art RF signal source.

Further, the power response of prior art RF signal sources istemperature dependent. For a given control input d.c. voltage level, itsoutput power level changes responsive to changes in temperature.Accordingly, the aforementioned calibration procedure must also beperformed at different temperatures to ensure that a given control inputd.c. voltage level can generate the appropriate desired output powerlevel at a given temperature.

To explore the difficulties experienced in the prior art, FIG. 3 shows aprior art RF signal source 300, including an RF power amplifier 302 anda power controller circuit 304. Responsive to a user-supplied controlsignal on a terminal 306, RF power amplifier 302 varies the power levelof the output signal on line 308 to cause RF signal source 300 to outputa desired power level.

Power controller circuit 304 represents the control circuit fordetecting the amount of power output by RF power amplifier 302 andcontrolling RF power amplifier 302, if necessary, to ensure that thedesired power output level is maintained for a given control signalvoltage level on terminal 306. In FIG. 3, a line coupler 310, also knownas a directional coupler, samples a fraction of the output power on line308 and provides a sensed signal on a line 312 to a power detector 314of power controller circuit 304. In a typical prior art RF signalsource, power detector 314 may be implemented by, for example, aSchottky barrier diode-based rectifier.

Power detector 314 rectifies the sensed signal on line 312 and outputs arectified signal on a line 316 to an error amplifier 318. Erroramplifier 318 compares the signals on line 316 and terminal 306 andoutputs an error signal on line 320 to control the amount of poweroutput by RF power amp 302. In the example of FIG. 3, error amplifier318 is implemented by an operational amplifier, which is arranged in aconventional integrating amplifier arrangement.

If the d.c. voltage level on line 316 (i.e., the output of powerdetector 314) is lower than that at terminal 306 (i.e., theuser-supplied control signal), error amplifier 318 will integrate upwardto increase the d.c. voltage level of the error signal on line 320. Thiscauses RF power amplifier 302 to increase its output on line 308, whichis sensed through line coupler 310 by power detector 314. RF poweramplifier 302 continues to increase its output until the d.c. voltagelevels on line 316 and terminal 306 are approximately equal. As is seen,power controller 304 increases the output of RF power amplifier 302whenever the d.c. voltage level of the user-supplied control signalincreases (i.e., power controller acts to increase the output powerlevel responsive to an increase in the d.c. level of the user-suppliedcontrol signal) or whenever the power level of the RF signal sourceoutput on line 308 drops below the level specified by the user-suppliedcontrol signal (i.e., power controller 304 acts to correct a drop in theoutput power level).

The typical power detector 314 is, however, nonlinear in its rectifyingcharacteristics over a wide dynamic range, e.g., over 30 dB of theoutput signal on line 308. In other words, the transfer characteristicsbetween the sensed signal on line 312 and the rectified signal on line316 is nonlinear for different output power levels on line 308. Thisnonlinearity affects the response of error amplifier 318, which in turncauses the error signal on line 320, which controls the RF poweramplifier 302, to be nonlinear. As a result, prior art power control 304results in, for different power levels output by RF power amplifier 302,a nonlinear relationship between the voltage level of the user-suppliedcontrol signal on terminal 306 and the power level output by the RFsignal source on line 308.

Since a wide dynamic range, i.e., a wide range of output power levels,is necessary in modem communication systems, this nonlinearity must becompensated for in the control algorithm that generates theuser-supplied control signal at terminal 306. The same nonlinearity alsocomplicates the aforementioned calibration procedure, typicallyrequiring the sampling of the output power levels on line 308 atdiscrete increments in the voltage level of the user-supplied controlsignal.

As mentioned earlier, the transfer characteristics of power detector 314is temperature dependent. At different temperatures, the rectifiedsignal on line 316 may vary for a given power level of the sensed signalon line 312. In the prior art, this temperature dependence must again beaccounted for in the control algorithm that generates the user-suppliedcontrol signal at terminal 306 and in the calibration procedure, i.e.,by sampling the output power level on line 308 at differenttemperatures. As can be seen, the prior art power control architecturerequires elaborate compensation and calibration procedures.

Alternative approaches in the prior art to linearize the power detectorover a wide dynamic range unfortunately give rise to other problems. Forexample, another well known prior art power controller architectureutilizes a logarithmic amplifier strip structure as power detector 314to achieve linear power detection over a wide dynamic range. However,the aforementioned logarithmic amplifier strip structure tends to bebandwidth-limited, i.e., it is unable to linearize power detection athigh frequencies. For example, it has been found that the logarithmicamplifier strip structure does not satisfactorily provide linear powerdetection at above 800 MHz over a 30-40 dB dynamic range at a powersupply consumption level that is appropriate for low power applications.

In view of the above, what is desired is an improved signal amplitudecontrol system that is more linear in its transfer characteristics andis more temperature independent. Furthermore, the improved signalamplitude control system preferably operates at a high bandwidth and ahigh dynamic range to accommodate the requirements of modem RF circuits,such as those employed in wireless communication systems.

SUMMARY OF THE INVENTION

The present invention relates, in one embodiment, to a power controllercircuit for generating an error signal from an alternating currentoutput signal of a power amplifier, a reference voltage level, and acontrol signal. The power controller circuit includes a power samplercircuit for sampling a portion of the output signal and outputting asensed signal, which is proportional to a power level of the outputsignal. There is further included a variable gain amplifier having afirst variable gain amplifier input and a second variable gain amplifierinput. The first variable gain amplifier input is coupled to the powersampler circuit for receiving the sensed signal while the secondvariable gain amplifier input receives the control signal. The variablegain amplifier functions to output a variable gain amplifier outputsignal having a substantially constant peak-to-peak voltage irrespectiveof a power level of the output signal of the power amplifier.

There is further provided a power detector circuit coupled to thevariable gain amplifier for receiving the variable gain amplifier outputsignal and for outputting a rectified signal. Additionally, there isprovided an error amplifier having a first error amplifier input andsecond error amplifier input. The first error amplifier input is coupledto the power detector circuit for receiving the rectified signal whilethe second error amplifier input is coupled to the reference voltagelevel, whereby the error amplifier outputs the error signal responsiveto a difference between the rectified signal and the reference voltagelevel.

In another embodiment, the present invention relates to a method forobtaining a rectified signal from a first alternating current signal.The method includes the step of inputing the first alternating currentsignal into a variable gain amplifier to obtain a second alternatingcurrent signal. In this embodiment the second alternating current signalhas a substantially constant peak-to-peak voltage irrespective of apower level of the first alternating current signal. The method furtherincludes the step of rectifying the second alternating current signal,using a power detector circuit, to obtain the rectified signal, wherebya direct current level of the rectified signal is substantiallyproportional to the power level of the first alternating currentirrespective of the power level of the first alternating current.

These and other advantages of the present invention will become apparentupon reading the following detailed descriptions and studying thevarious figures of the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustration showing an RF signal source, itscontrol input, and its variable output.

FIG. 2 is a plot of the RF signal source output power level (shown indBm) vs. the d.c. voltage level of its control input.

FIG. 3 shows a prior art RF signal source.

FIG. 4 shows, in accordance with one embodiment of the presentinvention, an RF signal source having a linearized power controller.

FIG. 5A shows, in accordance with another embodiment of the presentinvention, a power detector circuit, including the temperaturecompensation circuitry, that is suitable for use in the inventive powercontroller.

FIG. 5B shows, in accordance with another aspect of the presentinvention, a technique for compensating for thermal errors using thedetector cell of FIG. 5A.

FIG. 6 shows, in accordance with yet another embodiment of the presentinvention, a linearized power sensor circuit.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

As discussed above, FIG. 1 is a block diagram illustration showing an RFsignal source, its control input, and its variable amplitude-controlledoutput signal. FIG. 2 is a plot of the RF signal source output powerlevel (shown in dBm) vs. the d.c. voltage level of its control input.FIG. 3 shows a prior art RF signal source.

The invention relates, in one embodiment, to a power controller thatadvantageously provides linear power detection (i.e., detects the poweroutput by the RF power amplifier in a linear manner). Unlike prior artpower controller schemes, the power controller of the present embodimentremains linear over a wide range of RF amplifier output power, e.g., upto 40 dB, and at high frequencies, e.g. 800 megahertz and above, up to2.5 gigahertz.

FIG. 4 illustrates, in one embodiment, an RF signal source 400 thatadvantageously employs the inventive power controller 401 to sense andregulate the amount of power output by an RF power amplifier 402. RFpower amplifier 402 represents a voltage-controlled gain amplifier of aconventional design in which the power, i.e., the amplitude, of its a.c.output signal (herein "amplitude-controlled signal") on line 404 variesresponsive to the d.c. level of its control signal on line 406.

The amplitude-controlled signal on line 404 is sampled by a powersampler circuit 408, which outputs an alternating current (a.c.) sensedsignal on line 410, whose power is proportional to the power level ofthe amplitude-controlled signal on line 404. In one embodiment, powersampler circuit 408 represents the familiar directional coupler or linecoupler device for sampling a portion of the power level on line 404. Itis contemplated, however, that other known power sampler circuitdesigns, including transformer-based designs, may well be employed inpower sampler circuit 408.

The sensed signal on line 410 has substantially the same bandwidth anddynamic range as those of the signal it represents, i.e., theamplitude-controlled signal on line 404. If this sensed signal isrectified directly by a power detector circuit, as was done in the priorart by power detector 314 of prior art FIG. 3, the wide dynamic range ofthis sensed signal will result in a nonlinear transfer characteristicbetween the input and the output of the power detector. This is becausepower detectors are, as is well known, inherently nonlinear over a widedynamic range, e.g., above 10 dB.

To facilitate highly linear power detection over a wide dynamic range,e.g., above about 30 dB and up to about 40 dB in one embodiment, as wellas at high frequencies, e.g., above about 800 mega-hertz and up to about2.5 giga-hertz as in the case of wireless or cellular communicationsystems, the inventive power controller circuit advantageously sensesthe wide dynamic range output signal on line 404 but presents a signalhaving only a narrow dynamic range to the input of the power detector.With reference to FIG. 4, the sensed signal on line 410 is first"conditioned" by a variable gain amplifier (VGA) 412, which takes as itsinput the sensed signal on line 410 and either amplifies or attenuatesthe power of the sensed signal to output, responsive to a user-suppliedcontrol signal on line 414, a variable gain amplifier (VGA) outputsignal on line 416.

The aforementioned user-supplied control signal represents, in oneembodiment, the signal used for controlling the amount of power outputby RF power amplifier 402. This user-supplied control signal may begenerated by any number of conventional methods. In one embodiment, thisuser-supplied control signal is generated by a microcontroller, amicroprocessor, or digital logic circuitry through a digital-to-analogconverter (DAC) of a conventional design.

VGA 412 represents, in one embodiment, the well-known variable gainamplifier and is preferably selected from conventional designs thatpermit amplification (and concomitantly attenuation) over the highdynamic range and the high frequencies of interest. In one example, VGA412 may comprise multiple cascaded VGA stages to improve its dynamicrange and bandwidth.

Further, VGA 412 may be selected to have a gain control that is eitherlinear in voltage (i.e., its output voltage varies linearly with thevoltage level of its control signal on line 414) or, preferably, linearin dB (i.e., the output signal power in watts varies exponentially withthe voltage level of its control signal on line 414) for a high dynamicrange. The selection of an appropriate VGA 412 among the conventionaland known designs for a particular dynamic range and other power controlcharacteristics is well within the ability of those skilled in the art.

As is seen in FIG. 4, power sampling circuit 408, VGA 412, powerdetector 418, and error amplifier 424 form the feedback control loop ofRF signal source 400. When the control loop of RF signal source 400 isstable, VGA 412 will attenuate large sensed signals on line 410 oramplify small sensed signals on line 410 to maintain a relativelyconstant amplitude for its a.c. output signal on line 416.Advantageously, the VGA output signal on line 416 has a dynamic rangethat is substantially narrower than that of the sensed signal on line410.

The VGA output signal on line 416 is then input into a power detectorcircuit 418. Power detector circuit 418 represents a circuit of aconventional design for obtaining a d.c. output signal from an a.c.input signal. To function in a communication system, e.g., a wireless orcordless phone, the power detector circuit should be able to detectpower at the high operating frequencies of these systems. Suitable powerdetector circuits include those employing diodes, full wave rectifiers,or thermal transducers such as bolometers.

Because power detector 418 only has to rectify a signal having a narrowdynamic range, i.e., a relatively constant amplitude a.c. signal, thedynamic range-related nonlinearity is advantageously removed from thecontrol loop of RF signal source 400. Furthermore, the voltage gain ofVGA 412 may be chosen such that detection occurs near or at the optimumdetection point of power detector circuit 418, i.e., the point at whichthe detection characteristics of the power detector is most linear andthe bandwidth is maximized. In so doing, power detection accuracy isfurther improved. Without the fluctuations in the power level of itsinput signal, power detector circuit 418 performs power detection in amore linear manner, advantageously simplifying any correction that hasto be performed at that power range (due to, e.g., process variations inthe manufacture of the components of power controller 401) as well assimplifying the calibration of the resulting RF signal source 400.

The relatively constant amplitude of the VGA output signal on line 416produces a relatively constant d.c. signal at the output of the powerdetector circuit. This rectified signal (shown on line 420 of FIG. 4) isthen compared against a reference signal on a terminal 422 by an erroramplifier 424. In the embodiment of FIG. 4, error amplifier 424 isimplemented by an operational amplifier, which is configured in aconventional integrating amplifier arrangement. It is contemplated,however, that other conventional and well-known amplifying circuits forcomparing the difference between its input signals and outputting anerror signal may well be employed.

Responsive to the difference between its two inputs, error amplifier 424generates an error signal on line 406, which is then used as a controlsignal to set the gain of RF power amplifier 402. In this manner, aclosed loop is formed with power controller circuit 401 sensing, throughpower sampler circuit 408, the level of power output by RF poweramplifier 402 and adjusting that power level to correspond to the powerlevel set by the user-supplied control signal on line 414. The advantageof the present architecture lies, among others, in the fact that powerdetection by power detector circuit 418 is performed within a narrowdynamic range, thereby improving linearity while preserving both thehigh bandwidth and wide dynamic range.

Reference signal 422 is either a fixed d.c. voltage level, a temperaturecompensated d.c. voltage level, or a variable d.c. voltage signal thatis either internally or externally supplied. A variable reference signalmay be used to, for example, adjust power controller 401 to compensatefor process variations in the manufacture of the components of RF signalsource 400, e.g., in the gain error of RF power amplifier 402, in powerdetector 418, or in other components of the RF signal source, so that agiven user-supplied control signal on line 414 will generate the samelevel of output power on line 404. In this manner, absolute accuracyacross RF signal sources can be set.

To illustrate the operation of the inventive power control circuit,consider the situation where the user wishes to increase the power leveloutput by RF signal source 400. To do so, the d.c. voltage level of theuser-supplied control signal on line 414 is increased. Temporarily, thisdecreases the a.c. level, i.e., the peak-to-peak and RMS voltage level,of the VGA output signal on line 416. In turn, the rectified signal online 420 out of power detector circuit 418 is decreased. Error amplifier424, upon sensing the decreased voltage level on line 420 will cause anincrease in the voltage of the error signal on line 406, which in turnincreases the amount of power output by RF power amplifier 402.

This increased power level output by RF power amplifier 402 on line 404will be sensed by power sampler circuit 408, resulting in an increase inthe sensed signal on line 410. The power level on line 404 (andconcomitantly on line 410) will continue to increase until VGA 412 pullsthe VGA output signal on line 416 back up to its previous a.c. level tostabilize the feedback loop. At this point, the power output by RFsignal source 400 is stable and proportional to the d.c. voltage levelof the user-supplied control signal on line 414.

Conversely, a decrease in the voltage level of the user-supplied controlsignal on line 414 will result in a temporary increase in the a.c. levelof the VGA output signal on line 416. Correspondingly, the rectifiedsignal on line 420 is increased, resulting in a decrease in the voltagelevel of the error signal on line 406. Consequently, less power isoutput by RF power amplifier 402 and sensed by power sampler circuit408. The power level output by RF power amplifier 402 will continue todecrease until VGA 412 pulls the VGA output signal on line 416 back downto its previous a.c. level to stabilize the feedback loop. At thispoint, the power output by RF signal source 400 is again stable andproportional to the d.c. voltage level of the user-supplied controlsignal on line 414.

As can be seen from the foregoing, the present invention advantageouslyemploys one or more VGA stages to "absorb" the dynamic range of theoutput signal on line 404 and presents a more limited dynamic range topower detector circuit 418. Advantageously, the linearity of the powerdetection process is improved, with a corresponding improvement in thelinearity of the relationship between the voltage level of theuser-supplied control signal on line 414 and the level of power outputin dBm on line 404. If VGA 412 is chosen to be linear in voltage(instead of linear in dB), note that this linearity would appear in therelationship between the voltage level of the user-supplied inputcontrol signal on line 414 and the voltage level of the output signal online 404, which tends to increase detection accuracy, albeit at areduced dynamic range. In any case, this linear amplitude controlfeature permits simple calibration schemes to be employed,advantageously permitting the inventive RF signal source to becalibrated using only two calibration points (as discussed earlier inconnection with FIG. 2).

Most importantly, the power control circuit of FIG. 4 can operate at ahigh bandwidth and a wide dynamic range. For example, it is found thatthe inventive architecture is highly suitable for communication systemswhich specifies 30-40 dB of dynamic range at frequencies above 800megahertz up to 2.5 gigahertz, e.g., cordless phones, cellular phones,or PCS phones utilizing, for example, the GSM protocol.

In addition to the aforementioned benefits, it is found that the powercontrol architecture of the present invention can be realized onstandard high speed bipolar technology. It should be borne in mind atthis point that although high speed bipolar technology is preferred, itis contemplated that other technologies such as GaAs, CMOS, BiCMOS, maywell apply. Using standard technologies, it is possible to integratemost components of power controller 401 on a single integrated circuit(IC) chip, with an optional provision for externally coupling thecapacitor C1 of error amplifier 424. The reference signal at terminal422 may be internally provided or it may be externally specified ormodified through a pin on the IC. Compared to existing power controlschemes which utilize discrete components, the ability to functionallyintegrate the components of power controller 401 into a single ICadvantageously saves board space and component cost.

In another embodiment, an additional temperature compensation circuit ofa conventional design may be provided to correct for temperature-relatederrors in power detector circuit 418. In one example, a temperaturecompensation circuit is integrated into the design of a full waverectifier, i.e., fabricated on the same IC to implement power detectorcircuit 418.

FIG. 5A shows, in accordance with another embodiment of the presentinvention, a power detector circuit, including the temperaturecompensation circuitry, that is suitable for use in the inventive powercontroller. In FIG. 5A, a temperature-compensated logarithmic full-wavedetector is employed as a power detector. The detector biasing schemerelies on the inherent excellent matching of integrated bipolartransistors. Transistor pairs Q1/Q2 and Q7/Q8 are connected in the formof emitter followers that buffer the differential detector cell input,Vin. Transistors Q1 and Q2 have a relative area ratio of N:1, as dotransistors Q7 and Q8. These emitter followers are all biased at thesame current, Ip, via current sources I1, I2, I3, and I4. Under theseconditions, the voltage between emitters, shown in FIG. 5A as Vp, isindependent of the bias current Ip and is proportional to absolutetemperature (PTAT).

    Vp=ΔVbe=Vt 1n(N)                                     (Eq. 1)

This PTAT voltage Vp is used to bias the full-wave detector cell formedby transistors Q3-Q6. Transistors Q3 and Q4 have a relative area ratioof M:1, as do transistors Q5 and Q6. Each differential pair is biasedwith a current Ik. For accurate logarithmic operation, current Ik isusually set to be constant with temperature.

Transistor pair Q3 and Q4, which are connected between the emitters oftransistors Q2 and Q7, rectify positive input transitions. Transistorpair Q5 and Q6, which are connected between the emitters of transistorsQ1 and Q8, rectify negative transitions at input Vin. The detectoroutput is taken from the collectors of transistors Q3 and Q5 and isshown in FIG. 5A as current IR. The collectors of transistors Q4 and Q6may be tied to the supply voltage and are typically not used. The PTATbias voltage Vp, which is provided by the input emitter followers, isimposed across the differential pairs of the detector transistors.

For Vin=0:

    IR=+2IK/(M exp(Vp/Vt)+1)                                   (Eq. 2)

Substituting Eq. 1 into Eq. 2 and simplifying gives:

    IR=2IK/(M N+1) For Vin=0                                   (Eq. 3)

As seen in Eq. 3, the bias point is independent of temperature forVin=0. This result is due to the fact that the exact PTAT voltage Vp,which is derived from unequal area transistors biased with identicalcurrents, compensates for the temperature dependent bias required by thefull wave detector transistors.

The value of N and M may be chosen as appropriate depending on therequirements of an application. It has been found that M=N=3 is suitableand provides a bias ratio of 10:1. Larger M and N values yield largercurrent ratios but possibly at the cost of high speed operation sincetransistors Q3 and Q5 will be biased at very low currents where speed ispoor.

If Vin is included in the analysis, the dynamic current at the detectoroutput is then:

    IR(Vin)=IK/(1+M N exp(-Vin/Vt)+IK/(1+M N exp(Vin/Vt)       (Eq. 4)

The full-wave detector cell described above has a substantially constantslope w.r.t. temperature at the detector output as a function of inputpower (i.e., proportional to Vin²) presented to the input port over arange of about an octave. As indicated in Eq. 4, however, the detectorwill yield an output whose d.c. level will still vary with temperaturefor Vin not equal to zero.

FIG. 5B shows, in accordance with another aspect of the presentinvention, a technique for compensating for this thermal error in apower amplifier control system. The technique involves using a second,redundant full-wave detector cell 460 to buffer the fixed referencevoltage Vref. The output currents of detectors 460 and 462 of FIG. 5Bare converted to voltages using resistors R3 and R4, which havesubstantially similar resistance values.

A capacitor C3 is optionally added to the detector output of detector462 (i.e., the one coupled to the VGA) to pre-filter the rectifiedoutput current to an RMS d.c. level. The overall configuration providestemperature tracking to the actual error amplifier reference voltage tocreate a full-wave detector that is substantially temperaturecompensated under steady-state loop conditions. The nominal d.c. inputreference voltage, Vref, is preferably provided on chip. Provisions maybe added to allow the user to externally adjust the voltage of this d.c.input reference voltage, Vref, to calibrate the absolute accuracy of theentire power amplifier control system.

It should be borne in mind that the temperature compensated full-wavedetector architectures of FIGS. 5A and 5B represent only exemplary waysto implement power detection and that there exist other known circuitsfor implementing the aforesaid power detection. Consequently, thesepower detection implementations are included herein for illustrationpurposes and should not be construed as limiting the scope of theclaimed invention.

Likewise, conventional temperature compensation techniques may also beapplied to VGA 412 to correct for temperature related errors therein. Byway of example, temperature compensation techniques based on the use ofa proportional-to-absolute-temperature (PTAT) current, which compensatesfor temperature-related gain variations in bipolar transistors, may beemployed. Such temperature compensation circuitry may also be integratedinto the same IC chip to save board space and component cost. Note thatwhen such temperature compensation circuits are employed, the resultingRF signal source can be highly linear in terms of its amplitude controlover a high bandwidth and a wide dynamic range and is alsoadvantageously temperature stable.

In an alternative embodiment, the power controller of the presentembodiment may be configured as a high bandwidth power sensor to providean accurate and linear measure of the power output by an RF poweramplifier over a wide dynamic range. The power sensor output may be readby, for example, a microcontroller, a microprocessor, or other suitabledigital logic circuitry. Based on the level of power read, a controlsignal may then be provided to the RF power amplifier to modify itsoutput.

FIG. 6 shows one implementation of this power sensor embodiment. Inpower sensor 502, RF power amplifier 402, RF power sampler circuit 408,VGA 412, power detector circuit 418, reference terminal 422, and erroramplifier 424 are substantially similar to like components discussed andshown in FIG. 4. Note that no user-supplied control signal is input intoVGA 412 in FIG. 4. Instead, the control input into VGA 412 is coupled(via line 510) to the error signal out of error amplifier 424. The sameline 510 may also be used to couple the error signal to ananalog-to-digital converter (ADC) 512 to provide a digitalrepresentation of this error signal into, for example, a microcontroller514 (some microcontrollers may already have onboard an ADC, in whichcase ADC 512 may not be necessary). Responsive to this error signal,microcontroller 514 may then output a control signal on line 516 toadjust the output power level of RF power amplifier 402 to achieve adesired output power level.

It should be noted that the aforementioned advantages associated withthe power controller of FIG. 4 also applies to the power sensor of FIG.6. These advantages include: Highly linear power detection, temperatureindependent power detection, high bandwidth, wide dynamic range, and theability to integrate (with or without the temperature compensationcircuits) using standard technologies.

While this invention has been described in terms of several preferredembodiments, there are alterations, permutations, and equivalents whichfall within the scope of this invention. It should also be noted thatthere are many alternative ways of implementing the methods andapparatuses of the present invention. It is therefore intended that thefollowing appended claims be interpreted as including all suchalterations, permutations, and equivalents as fall within the truespirit and scope of the present invention.

What is claimed is:
 1. A power controller circuit for generating anerror signal from an alternating current output signal of a poweramplifier, a reference voltage level, and a control signal, comprising:apower sampler circuit for sampling a portion of said output signal andoutputting a sensed signal, said sensed signal being proportional to apower level of said output signal; a variable gain amplifier having afirst variable gain amplifier input and a second variable gain amplifierinput, said first variable gain amplifier input being coupled to saidpower sampler circuit for receiving said sensed signal, said secondvariable gain amplifier input receiving said control signal, foroutputting a variable gain amplifier output signal having asubstantially constant peak-to-peak voltage irrespective of a powerlevel of said output signal; a power detector circuit coupled to saidvariable gain amplifier for receiving said variable gain amplifieroutput signal, said power detector circuit outputting a rectifiedsignal; an error amplifier having a first error amplifier input andsecond error amplifier input, said first error amplifier input beingcoupled to said power detector circuit for receiving said rectifiedsignal, said second error amplifier input being coupled to saidreference voltage level, whereby said error amplifier outputs said errorsignal responsive to a difference between said rectified signal and saidreference voltage level; and whereby said variable gain amplifier isconfigurable to have either a substantially linear in dB gain control ora substantially linear in voltage gain control.
 2. The power controllercircuit of claim 1 wherein said power detector circuit further comprisesa first temperature compensation circuit to render a transfercharacteristic of said variable gain amplifier substantially independentof temperature variations.
 3. The power controller circuit of claim 2wherein said error amplifier circuit further comprises a secondtemperature compensation circuitry to render a transfer characteristicof said power detector circuit substantially independent of temperaturevariations.
 4. The power controller circuit of claim 1 wherein saidfirst alternating current has a dynamic range of about 40 dB.
 5. Thepower controller circuit of claim 4 wherein a frequency of said firstalternating current is above about 800 mega-hertz.
 6. The powercontroller circuit of claim 5 wherein a frequency of said firstalternating current is up to about 2.5 giga-hertz.
 7. The powercontroller circuit of claim 4 wherein a frequency of said firstalternating current is up to about 2.5 giga-hertz.
 8. A method forobtaining a rectified signal from a first alternating current signal foruse with a power amplifier having an output signal comprising:inputtinga control signal and said first alternating current signal into avariable gain amplifier to obtain a second alternating current signal,said second alternating current signal having a substantially constantpeak-to-peak voltage irrespective of a power level of said firstalternating current signal; rectifying said second alternating currentsignal, using a power detector circuit, to obtain said rectified signal,whereby a direct current level of said rectified signal is substantiallyproportional to said power level of said first alternating currentirrespective of said power level of said first alternating current; andwhereby said variable gain amplifier is configurable to have either asubstantially linear in dB gain control or a substantially linear involtage gain control.
 9. The method of claim 8 wherein said powerdetector comprises a first temperature compensation circuit to render atransfer characteristic of said variable gain amplifier substantiallyindependent of temperature variations.
 10. The method of claim 9 whereinsaid error amplifier circuit further comprises temperature compensationcircuitry to render a transfer characteristic of said power detectorcircuit substantially independent of temperature variations.
 11. Themethod of claim 8 wherein said first alternating current has a dynamicrange of about 40 dB.
 12. The method of claim 11 wherein a frequency ofsaid first alternating current is above about 800 mega-hertz.
 13. Themethod of claim 12 wherein a frequency of said first alternating currentis up to about 2.5 giga-hertz.
 14. The method of claim 8 wherein saidfirst alternating current has a dynamic range of about 30 dB to about 40dB.